International Journal of Scientific & Engineering Research, Volume 5, Issue 2, February-2014 1524
ISSN 2229-5518
Analysis & Designing of RF transistor as an
Amplifier with its parametric limitations
Asst.prof Purvi Zaveri
Babaria Institute of Engineering and Technology,Varnama-Baroda(Gujarat)-India
Email:purvizaveri@yahoo.co.uk
Abstract— Amplifier play a critical role in acommunication system.They are used to increase the voltage,current,and /or power of a signal, in both transmitters and receivers.RF amplifier is used at the front end of the receivers.Important properties of amplifiers include gain,input and output impedances measured using S-parameters,noise figure,stability,bias networks,interface with other circuit (ports).W hen designing a linear narrow band amplifier,the ggoal is to produce a design that is well matched at the design frequency and has a good gain and low noise at that frequency.The S-parrameters are a function of bias condition,so the result using a set of S-parameters are only valid at the bias conditions at which the S-parameters were measured or simulated.In this paper,using bilatersl approach RF Amplifier is designed using RF transistor BFP 640(product of Infineon Technology).The BFP 640 gives maximum gain with higher noise figure by keeping VSW Rin as constant whereas,plot of VSW R (input/output) variations are observed separately.
Index Terms— 4
—————————— ——————————
In Communication System LNA (low noise amplifier) pro- vides first level of amplification of the signal received at the system’s antenna as shown in figure 1. It plays an undisputed
important role in the receiver design. Its main function is to amplify extremely low signals without adding noise, thus pre- serving required signal to noise ratio of the extremely low power levels and for large signal levels, it amplifies the re- ceived signal without introducing any distortions, hence elim- inating channel interference. The most important design con- siderations in a RF Amplfier design are stability, power gain, bandwidth, noise figure, VSWR and DC requirement.
Fig. 1. Communication System [14]
Selection of proper transitor by examining datasheets.Checking the conditional stability of transis- tor.Biasing is selected depending on the application like low power,low noise,high power,linearity,etc.Different techniques are applied to optimize different parameters.Two parameters can not be optimized simultaneously.Matching circuits that provide optimum performance in a microwave amplifier can be easily and quickly designed using smith chart.Lumped components are placed with distributed transmission lines.In
the design of a LNA, the transistor should have minimum in- trinsic noise.Noise sources present in a transistor should be properly modeled.
good input and output matching and unconditional stability
at the lowest possible current draw from the amplifier. All
these parameters are interdependent and do not always work
in each other’s favour. Carefully selecting a transistor and un-
derstanding parameter limitations can meet most of these
conditions. Low noise figure and good input match can be obtained using feedback arrangements. Unconditional stabil- ity will always require a certain gain reduction because of ei- ther shunt or series resistive loading of the collector. There is a
major limitation to observe variations in VSWR parameters. It is not possible to plot input and output VSWR circles simulta- neously. A Plot of (Input / Output) VSWR shows that one of the network is matched properly and it is considered as refer- ence to observe variation in the other side of the network.
The stability of an amplifier, or its resistance to oscillate, is a very important consideration in a design and can be deter- mined from the S parameters, the matching networks, and the terminations. Instabilities are primarily caused by three phe- nomena (1) Internal feedback of the transistor, (2) External feedback around the transistor caused by external circuit, or (3) Excess of gain at frequencies outside the band of operation. To determine the stability of a device, calculate the Rollett’s stability factor (K) using a set of S-parameters given for the device at the frequency of operation. K and |Δ| give us an indication to whether a device is likely to oscillate or not or
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International Journal of Scientific & Engineering Research, Volume 5, Issue 2, February-2014 1525
ISSN 2229-5518
whether it is conditionally/unconditionally stable. The pa- rameter must satisfy K > 1 and |Δ| < 1 for a transistor to be unconditionally stable.
In terms of reflection coefficients, the conditions for uncondi- tional stability at a given frequency are
|ΓL |<1, |ΓS |<1 ∆=S11 S22 -S12 S21 (1)
1. Input matching network: It minimizes the noise influence. It is employed to match the input of the transistor to the source to provide a noise figure F as close to Fmin as possible over the amplifier band width.
2. Output matching network: It maximizes power
handling capabilities. It is required to match the
| Γin
| Γ
|=| S11 − ΓL ∆ |< 1
1 − S22ΓL
|=| S22 − ΓS ∆ |< 1
(2)
output of transistor to the load to provide the highest possible gain. There are two ways to obtain desired gain (1) Unilateral case (2) Bilateral case.
out
1− | S
1 − S11ΓS
|2 − | S |2
+ | ∆ |2
(3)
performance, the influence of the transistor’s feedback is neglected. ( S12 ≈ 0 )
k = 11 22 > 1
2 | S12 || S21 |
(4)
overcome the error introduced in unilateral design where feedback is not considered. ( S12 ≈ 0 )
Thus, the purpose of a matching network can be defined as a
The unconditionally stable conditions can be represented graphically also. The graphical analysis is especially useful in the analysis of potentially unstable transistors. The active de- vice would be showing unconditional stability, as there is no intersection of the stability circles on the Smith Chart as shown in figure 2.Thus, the device will be stable for all possible matches on the input or output of the active device. Figure 3 shows the second case where S11 or S22 > 1 the origin is the part of the unstable region. When S11 or S22 <1, the origin is part of the stable region [8].
transformation to convert given impedance value to another more suitable value. Figure 4 illustrates a typical situation in which a transistor, in order to deliver maximum power to the Zo = 50 Ω load, must have the terminations ZS and ZL .
Fig. 4. Block diagram of Microwave amplifier [8]
Fig. 2. Stability circles for unconditional stability[14]
Fig. 3. Area of Stability circles for conditional stability[14]
The network with strip lines and stub sections are more suita- ble for operational frequencies exceeding 1 GHz. With increas- ing frequency and corresponding reduced wavelength, the influence of parasitic in the discrete elements becomes more noticeable and taking their effect complicating the component value computations. As an alternative to lumped elements, distributed components are widely used when the wavelength becomes sufficiently small compared with the characteristic circuit component length. Even, it is easy to fabricate in micro strip or strip line form with a microwave fabrication aspect, since lumped elements are not required. To minimize the size of the circuit board, try to employ the shortest possible trans- mission line segments.
Particularly, in this paper for designing matching network, single stub in shunt form, balanced stub and impedance trans- forming properties of transmission lines are used. The shunt component can be open/short-circuit stub lines depending on the impedance requirements. The single stub in shunt form is shown in figure 5.
The impedance matching networks can be designed either
mathematically or graphically with the aid of Smith Chart.
Here, graphical approach of ZY Smith Chart is used in the
design of matching networks.
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as matched source and load reflection coefficients respective- ly, ΓMS and ΓML .
The maximum transducer gain GT is given by
ZOL,lL
(1− | Γ
|2 ) | S
|2 (1− | Γ
|2 )
G = ML 21 MS
T ,max
| (1 − S Γ
)(1 − S Γ
) − S S Γ Γ |2
11 MS
22 ML
21 12
ML MS
(5)
Maximum gain is given by
Zin
ZOS,lS
ZL
Open /
GT max =
S21 K −
S12
K 2 −1)
(6)
short - circuit
Fig. 5. Single Stub Matching Network [8]
ST2
Length in λ
TL1
Length in λ
Zin ST1
Length in λ
L
Similarly, Operating power gain GP can also be solved and it
can be represented in circle equations to plot operating power
gain circle [11].
For RF amplifiers, the need for signal amplification at low noise level becomes an essential system requirement. As de- signing a low noise amplifier competes with factors as stability and gain. For instance, a minimum noise performance at max- imum gain cannot be obtained. It is necessary to develop a method that allows us to display the influence of noise as part of the smith chart to conduct comparisons and observe trade- offs between gain and stability.
The noise figure circle can be represented as
Fig. 6.Balanced Stub and Matching Network [8]
In practical realizations single sided unbalanced stubs are re-
F = Fmin
+ 4Rn
2
ΓS − Γopt
2 2
placed by the balanced stub along the series transmission line
(TL1) to minimize transition interaction between the shunt
Zo (1 − ΓS
) 1 + Γopt
(7)
stubs and the series transmission lines as shown in figure
6.Two parallel shunt stubs ST1 and ST2 must provide the
same admittance as the single stub. Therefore, the admittance
of each side of the balanced stub must be equal to half of the
total admittance. It is important to note that the length of the
shunt stubs is not equal to the total length of the balanced
stubs.
Impedance transforming properties of transmission line uses
micro strip lines with different characteristic impedances. A
micro strip line can be used as a series transmission line, as an
open circuited stub or as a short circuited stub. A series micro
strip line together with a short-or open circuited shunt stub can transform a 50 Ω resistor into any value of impedance. Also, a quarter wave micro strip line can be used to change a
50 Ω resistor to any value of resistance. This line, together with a short-or open circuited shunt stub, can be used to transform
50 Ω to any value of impedance [8].
There are three basic important parameters for calculations of power gain. They can be defined as transducer power gain, operating power gain and available power gain. They depend upon the source, load, input, output reflection co-efficient and S-parameters. Bilateral design is considered as simultaneous conjugate match. When S12 ≠0, and the unilateral assumption cannot be made. Solving input and output reflection equations to obtain the simultaneous conjugate match. They are known
In above equation the quantities Fmin (minimum noise figure), Rn (noise resistance of the device), Γopt (optimum source reflec- tion coefficient) are known. Adjust ΓS such that, ΓS = Γopt so that the lowest possible noise figure is achieved, F=Fmin . The above equation is in the form which on the right hand side suggests the form of a circle equation. It can be solved to plot the noise figure circles. In a design there is always a difference between the designed noise figure and the measured noise figure of the final amplifier. This occurs because of the loss associated with the matching elements and the transistor noise figure variations from unit to unit [11].
The amplifier has to stay below a specified VSWR as meas- ured at the input or output port of the amplifier. Typical val- ues ranges between 1.5 ≤ VSWR ≤ 2.5. The purpose of match- ing networks is to reduce the VSWR at the transistor. The complication arises from the fact that the input VSWR (VSWRIMN) is determined at the input matching network (IMN), which in turn is affected by the active device, and, through feedback, by the output matching network (OMN). Conversely, the output VSWR (VSWROMN) is determined by the OMN and, again through feedback, by the IMN. Figure 7 shows the system configuration for input and output VSWR.
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ISSN 2229-5518
K | Δ | GT (dB) | Actual NF( dB) | VSWRi n | VSWRo ut |
1.0166 | 0.5113 | 19.13 19 18.5 | 2.7911 2.7113 2.76 | 1 1 1 | 1.970 1.2254 1.608 |
Fig. 7. System configuration for Input and output VSW R [8]
Below equations can be solved in form of circle equations to plot input/output VSWR circles. Under bilateral matching the input and output reflection coefficients are the functions of source and load reflection coefficients (ΓS , ΓL ).Therefore, the input and output VSWR circles cannot be plotted simultane- ously, but rather have to be considered one at a time in the iterative process of adjusting Γ S and ΓL [11].
k = stability factor, Δ = stability factor, G T = Transducer gain; NF= Noise fig- ure, VSWR =Voltage standing wave ratio for input and output matching net- work
TABLE-2
Calculation of Reflection Coefficients for simultaneous conju-
gate match for maximum gain 19.13 dB
VSWRIMN
VSWROMN
1 + Γ
= IMN
1 − ΓIMN
1 + Γ
= OMN
1 − ΓOMN
(8) (9)
Гms= reflection coefficient for matching source, Гml=reflection coeffiecient for matching load, y ms =admittance for matching source, y ml =admittance for match- ing load
An important building block in any RF circuit is the active or passive dc biasing network. The purpose of a good dc biasing is to provide the appropriate quiescent point for the active devices under specified operating conditions and maintain a constant setting irrespective of transistor parameter variations and temperature fluctuations. The biasing not only sets the DC operating conditions but must also ensure isolation of the RF signal through the use of radio frequency chokes (RFC) and blocking capacitors. In this design, the dc biasing network with a bypassed emitter resistor is employed [11].
MATLAB Simulation Results for Bilateral Design
Vce = 1.5 V , Ic = 20 mA , f = 2.4 GHz, Gain=19.13 dB
r = 0.10 Ω , F =1.03 , Γ = 0.15∠ −131°
Fig. 8. Stability Circles
Note: Gain circles show the variation of radius ΓL . As gain increases ΓL shrinks to zero as shown in figure 9, 10and 11.
Fig. 9. Operating Power Gain 18.5 dB
TABLE-1 1
Stability, Gain, Noise Fiure and VSWR Results
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Fig. 10. Operating Power Gain 19 dB
Fig. 14. For VSW RIN=2.5 , NF=1.5,and De- sired Gain 18.5 dB
Fig. 11. Operating Power Gain 19.13 dB
Note: Parametric variations of gain, noise figure and VSWR in graphical form.
Fig. 12. For VSW RIN=1.5 , NF=1.4,and Desired
Gain 18 dB
Fig. 15. Input and Output VSW R plot VSW Rin=1.5 and VSW Rout=1.49
Fig.16. Input and Output VSW R plot VSW Rin=1.8 and VSW Rout(min)=1.29 and VSW Rout(max)=2.03
Fig.13.ForVSW RIN=1.8,NF=1.8,andDesired
Gain 19 dB
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Fig.17.Input and Output VSW R plot
VSW Rin=2.5 and VSW Rout=1.15
Fig. 19. Amplifier schematic of BPF 640 (balanced stub)
Vcc=5 V
75 Ω
2.33 KΩ
RFC
C
bypass capacitor Ce is typically 0.01 µF disk capacitors. The
50 Ω 24.69 Ω
RFC
B
CB
BFP 640
129 Ω
RFC are typically made of 2 or 3 turns of No.36 enameled wire
on 0.1 in. air core. The chip capacitors CA are coupling capaci-
tors having typical values 200 to 1000 pF. The bypass capaci- tors CB are chip capacitors having typical values of 50 to 500
pF. The typical value of hFE is 100. The Stability factors are Si =
CA λ/4
15.625Ω
AC 3λ/8
RFC
3.5 KΩ
CB
100 Ω Ce
3λ/8
λ/4
142 Ω
50 Ω
7.769, ShFE = 9.23*10-6, SVBE = -9.23.Matching networks are de- signed with the help of smith chart for maximum gain 19.13 dB, with higher noise figure and moderate VSWR effect.The main limitation of this design is to have minimum noise figure and moderate VSWR at the cost of reduction in gain factor. As the gain changes, stub length will also be affected.
To overcome this problem, one can design the circuit with variable stub length. This helps to tune the circuit as per the requirement of amplifier’s application, like higher gain or low noise figure or low reflections.
BFP 640 RF transistor provides unconditional stability at 5-6
GHz as compared to 2.4 GHz.
.Several applications are mentioned like 2.4 GHz Wi-Fi wire- less LAN and WiMax front end, Global Navigation satellite system (GNSS) with RF discrete device at 1.5 GHz, Satellite
Fig. 20. Amplifier schematic of BPF 640 (single stub using different characteristic impedance)
Fig. 21. Amplifier schematic of BPF 640 (balanced stub us- ing different characteristic impedance)
digital multimedia broadcasting at 2.6 GHz. 8 CONCLUSION
Stability, gain, noise figure and VSWR performances are ob- tained using MATLAB. The matching circuits are calculated using Smith Chart. Three different networks are designed us-
Vcc=5 V
2.33 KΩ
RFC CB
50 Ω Tx Line 0.088 λ
CA
75 Ω
RFC
ing single stub, balanced stub and impedance transforming properties of transmission lines. The DC bias circuit is de- signed with the help of voltage divider configuration.
CB
CA
Tx Line 0.135λ MATLAB result shows that moderate noise figure can be ob-
BFP 640
RFC CB
AC
Short-circuit stub 0.0658 λ
50 Ω
tained for maximum gain or vice-versa, but it is difficult to
Open circuit stub 0.18 λ
3.5 KΩ
175 Ω Ce
realize the VSWR effect. VSWR plot shows that by keeping
one parameter constant, other parameter variations can be
realized. Certain mismatches are required at input side to have
Fig. 18. Amplifier schematic of BPF 640 (single stub)
minimum reflections at output side. That results in higher in-
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International Journal of Scientific & Engineering Research, Volume 5, Issue 2, February-2014 1530
ISSN 2229-5518
put VSWR. Thus, higher gain with minimum VSWR at output side can be obtained at the cost of higher input VSWR.
Some errors are likely to exist in the final implementation of the design resulting from parameter variations, stray capaci- tances and other random causes. ADS/HFSS Softwares are required for accurate analysis.
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